FAN5250 データシートの表示（PDF） - Fairchild Semiconductor

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FAN5250

Mobile Processor Core-Voltage Regulator

Fairchild Semiconductor 

FAN5250

Output Capacitor Selection

The output capacitor serves two major functions in a switch-

ing power supply. Along with the inductor it filters the

sequence of pulses produced by the switcher, and it supplies

the load transient currents. The filtering requirements are a

function of the switching frequency and the ripple current

allowed, and are usually easy to satisfy in high frequency

converters.

The load transient requirements are a function of the slew

rate (di/dt) and the magnitude of the transient load current.

Modern microprocessors produce transient load rates in

excess of 10A/µs. High frequency ceramic capacitors placed

beneath the processor socket initially supply the transient

and reduce the slew rate seen by the bulk capacitors. The

bulk capacitor values are generally determined by the total

allowable ESR rather than actual capacitance requirements.

High frequency decoupling capacitors should be placed as

close to the processor power pins as physically possible.

Consult with the processor manufacturer for specific decou-

pling requirements. Use only specialized low-ESR electro-

lytic capacitors intended for switching-regulator applications

for the bulk capacitors. The bulk capacitor’s ESR will deter-

mine the output ripple voltage and the initial voltage drop

after a transient. In most cases, multiple electrolytic capaci-

tors of small case size perform better than a single large case

capacitor.

Power MOSFET Selection

For the example in the following discussion, we will be

selecting components for:

VIN from 5V to 20V

VOUT = 1.2V @ ILOAD(MAX) = 7A

The FAN5250 converter's output voltage is very low with

respect to the input voltage, therefore the Lower MOSFET

(Q2) is conducting the full load current for most of the cycle.

Therefore, Q2 should be selected to be a MOSFET with low

RDS(ON) to minimize conduction losses.

In contrast, Q1 is on for a maximum of 20% (when VIN =

5V) of the cycle, and its conduction loss will have less of an

impact. Q1, however, sees most of the switching losses, so

Q1’s primary selection criteria should be gate charge

(QG(SW)).

High-Side Losses:

VDS

C ISS

C RSS

C ISS

ID

VGS

QGS

QGD

VSP

VTH

QG(SW)

t1

t2

t3

CISS = CGS || CGD

4.5V

t4

t5

Figure 14. Switching Losses and QG

5V

RD

VIN

19 HDRV

20 SW

CGD

RGATE

G

CGS

Figure 15. Drive Equivalent Circuit

Assuming switching losses are about the same for both the

rising edge and falling edge, Q1’s switching losses, as can be

seen by Figure 14, are given by:

PUPPER = PSW + PCOND

PSW

=





V-----D---S--2---×-----I--L-

×

2

×

tS

FSW

PCOND = V---V--O---I-UN---T-- × IO2 UT × RDS(ON)

(14a)

(14b)

(14c)

Where RDS(ON) is @TJ(MAX) and tS is the switching period

(rise or fall time) and is predominantly the sum of t2, t3

(Figure 14), a function of the impedance of the driver and the

QG(SW) of the MOSFET. Since most of tS occurs when VGS

= VSP we can use a constant current assumption for the

driver to simplify the calculation of tS:

tS = I-Q-D----GR----(I--VS---EW---R--) ≈ ---R----------D--------R--V-----I---DV-Q-------E--D-G---R----(----–S--+------WV----R----S-)---G----P----A--------T------E------

(15)

14

REV. 1.1.6 3/12/03

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